VCM driver and PWM amplifier

ABSTRACT

The present invention provides a VCM driver realizing low power consumption and high accuracy and a PWM amplifier compensating a dead time distortion. A phase compensator, a ΔS modulator receiving an output signal of the phase compensator and converting the output signal to a control code of predetermined bits, a PWM modulator receiving the control code to produce a PWM signal, and an output circuit receiving the PWM signal to drive a voice coil constitute a forward path. A sense amplifier sensing a current of the voice coil, an ADC receiving an output signal of the sense amplifier, a low-pass filter receiving an output signal of the ADC, and a decimation filter receiving an output signal of the low-pass filter constitute a feedback path. An output signal of the decimation filter is fed back to the input side of the phase compensator to form a major feedback loop having a first-order characteristic loop gain. An output signal of the decimation filter is fed back to the output side of the phase compensator to form a minor feedback loop having a loop gain which is flat until a target band frequency when viewed from the output side of the phase compensator.

CROSS-REFERENCE TO RELATED APPLICATION

The present application claims priority from Japanese patent applicationNo. 2007-243662 filed on Sep. 20, 2007, the content of which is herebyincorporated by reference into this application.

BACKGROUND OF THE INVENTION

The present invention relates to a voice coil motor (VCM) driver and apulse width modulation (PWM) amplifier, and in particular to atechnology effectively applied to a VCM driver and a PWM amplifier usedfor, for example, a hard disk memory device.

In a hard disk drive device, seek-time needed for movement betweentracks tends to be shortened from the viewpoint of high speed access. Asa result of this, the drive current of a voice coil motor (VCM) hasincreased and heat generation at a seek operation has become a problem.In order to solve this heat generation problem, the power consumption isreduced by pulse width modulation (PWM) drive only in the seek period oftime when accurate positioning control is not required but powerconsumption is large. On the other hand, at the time of track followingwhen the magnetic head is allowed to follow a required track for readand write, high-precision control is required for accurate tracking andthe influence of a noise caused by PWM drive should be avoided, so thatlinear drive is used instead of PWM drive. An example of a VCM driverusing such a PWM/linear combination method is disclosed in JapanesePatent Laid-Open No. 2002-184137 (Patent Document 1). Furthermore, anexample of a VCM driver entirely using PWM drive is disclosed inJapanese Patent Laid-Open No. 2005-304095 (Patent Document 2) andJapanese Patent Laid-Open No. 2005-304096 (Patent Document 3). All ofthem have been proposed by the present inventors.

SUMMARY OF THE INVENTION

The recording density of a hard disk tends to be further increased byadopting a vertical recording method or the like in order to increasethe storage capacity per unit area of the hard disk. When the recordingdensity is increased like this, the track interval is reduced and theaccurate tracking thus becomes difficult in the PWM method. For example,also in linear operation by a PWM/linear combination method shown inPatent Document 1, a digital-to-analog converter (DAC) converting adrive current value set at a digital value to an analog value isprovided and a VCM driver is driven with the output value of the DAC. Atthat time, also in a liner method shown in Patent Document 1, aquantization noise, a 1/f noise, or a white noise which arises atdigital-to-analog conversion becomes a large obstacle at a trackfollowing time when the magnetic head is allowed to follow a requiredtrack for read and write.

In the VCM driver of Patent Document 2 or 3, PWM drive is entirelyadopted by devising methods of (1) using a current amplifier with aphase compensator realized by a digital filter, (2) reducing aquantization error caused by PWM modulation of an output by ΔSmodulation, (3) reducing a switching error by measuring the delay timeand transition time of a switching waveform of an output stage, and (4)reducing a PWM modulation error of an output stage by measuring thepower supply voltage of the output stage with the ADC. In this case, anoise generated by the DAC and a noise caused by the phase compensatorof the VCM driver are reduced. However, a high recording densitycorresponding to a vertical recording method or the like as describedabove causes the following problems.

In the PWM amplifier of Patent Document 2, a control dead band arisesnear a zero-crossing point of a coil current by a dead time of theoutput switching circuit, thus causing a large noise (occurrence of azero-crossing distortion). As a result of this, accurate trackingbecomes impossible and the output distortion becomes large for the PWMamplifier. Furthermore, an ADC is necessary to correct an output error(PWM error) to a power supply variation, thereby increasing the cost.The update rate of an input signal is not synchronized with the updaterate of the output-side ΔS modulator and there is little frequencydifference between them, so that the generation of a beat noise isfeared. A control band is restricted by a delay caused by an LPF ordecimation filter subsequent to a ΔS ADC, so that it is difficult toachieve a control band desired by a hard disk drive (HDD) using avertical recording method.

An object of the present invention is to provide a VCM driver realizinglow power consumption and high accuracy. Another object of the presentinvention is to provide a PWM amplifier reducing a dead time distortionand suitable for a VCM driver and the like. The above and furtherobjects and novel features of the present invention will be apparentfrom the following description of this specification and theaccompanying drawings.

An embodiment of the present invention is as follows. A VCM driverproduces a drive current of a voice coil motor for position control of amagnetic head with a forward path and a feedback path. The forward pathincludes a digital operation type phase compensator receiving an inputsignal, a ΔS modulator receiving an output signal of the phasecompensator and converting the output signal to a control code ofpredetermined bits, a PWM modulator receiving the control code convertedby the ΔS modulator to produce a PWM signal, and an output circuitreceiving the PWM signal produced by the PWM modulator to drive a voicecoil. The feedback path includes a sense amplifier sensing a currentflowing in the voice coil, an ADC receiving an output signal of thesense amplifier, a first low-pass filter receiving an output signal ofthe ADC, and a first decimation filter receiving an output signal of thefirst low-pass filter. An output signal of the decimation filter is fedback to the input side of the phase compensator to form a major feedbackloop having a first-order characteristic loop gain. An output signal ofthe decimation filter is fed back to the output side of the phasecompensator to form a minor feedback loop having a loop gain which isflat until a target band frequency when viewed from the output side ofthe phase compensator. Another embodiment of the present invention is asfollows. A PWM drive circuit receives a PWM input signal to produce afirst drive pulse and a second drive pulse each having a dead time. Afirst output element receives the first drive pulse to output a firstoutput voltage to an output terminal. A second output element receivesthe second drive pulse to output a second output voltage to the outputterminal. A dead time compensation circuit detects an error pulsebetween the PWM input signal and an output signal produced by adifference potential between the first output element and the secondoutput element corresponding to the PWM input signal and adds the errorpulse to a PWM input signal to be input next.

A second-order error suppression characteristic can be obtained at adrive end of a VCM by a negative feedback effect of the minor feedbackloop and a negative feedback effect of the major feedback loop, so thatboth of an error and a zero-crossing distortion at an output stagecaused by a power supply variation are reduced and an accurate trackingoperation becomes possible. A zero-crossing (dead time) compensationcircuit makes accurate tracking possible and is able to improve anoutput distortion as a PWM amplifier.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of an embodiment of a VCM driver according tothe present invention;

FIGS. 2(A) and 2(B) show loop gain characteristic diagrams forexplanation of the present invention;

FIG. 3 is an error suppression characteristic diagram for explanation ofthe present invention; FIG. 4 is a block diagram of an embodiment of azero-crossing compensation circuit according to the present invention;FIG. 5 is a block diagram of an embodiment of an expanded ΔΣ modulator;

FIG. 6 is a block diagram of another embodiment of an expanded ΔΣmodulator;

FIG. 7 is a simulation waveform diagram corresponding to a VCM driver ofFIG. 14;

FIG. 8 is a simulation waveform diagram in the case that the minor loopin FIG. 1 is provided;

FIG. 9 is a simulation waveform diagram in the case that thezero-crossing compensation circuit in FIG. 1 is provided;

FIG. 10 is a simulation waveform diagram in the case that the minor loopand the zero-crossing compensation circuit in FIG. 1 are provided;

FIG. 11 is a block diagram of another embodiment of a VCM driveraccording to the present invention;

FIG. 12 is a block diagram of another embodiment of a VCM driveraccording to the present invention;

FIG. 13 is a schematic configuration diagram of an embodiment of amagnetic disk device to which the present invention is applied;

FIG. 14 is a block diagram of a VCM driver using a PWM method examinedby the present inventors;

FIGS. 15(A) and 15(B) illustrate an output circuit used in the presentinvention;

FIG. 16 is a transmission characteristic diagram of an output circuitfor explanation of the present invention; and

FIG. 17 is a characteristic diagram showing the relation between the PWMfrequency and the quantization error for explanation of the presentinvention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1 is a block diagram of an embodiment of a VCM driver according tothe present invention. The VCM driver of FIG. 1 has a forward path forsupplying a drive current to a voice coil VCM and a feedback path forsensing the drive current of the voice coil VCM. The forward pathincludes a phase compensator, an expanded ΔΣ modulator, MOD, PA, and NA.The feedback path includes Ksens, ADC, LPF1, DFL, and LLFL. The MOD is aPWM modulator. The PA and NA are output circuits. The Ksens is a senseamplifier. The LPF1 is a low-pass filter. The DFL is a decimationfilter. The LLFL is a lag-lead filter.

The forward path is as follows. An input signal IN is a drive currentcommand, for instance, a 16-bit digital signal. The input signal IN isproduced by a controller including a microcomputer described later, andis input to the forward path through a serial input/output port notshown in the figure. The update rate (frequency) of the input signal INis the order of 50 kHz in a present HDD, while being expected to becomehigher like 100 kHz in future.

An output of error comparison between the input signal IN (drive currentcommand) and the detection value Ivcmdet of a VCM current which haspassed through the feedback path is input to the phase compensator (PItype). An output signal of the PI type phase compensator is input to theexpanded ΔΣ modulator. The expanded ΔΣ modulator converts thephase-compensated error comparison output to a control codecorresponding to PWM. The PWM modulator MOD receives the control code toproduce a PWM pulse. The output circuits PA and NA constitute a bridgecircuit and PWM-drives the voice coil VCM coupled to them through theoutput terminals VCMP and VCMN. In other words, the expanded ΔΣmodulator converts the phase-compensated current value to a control codesignal of a predetermined number of bits (e.g. 7 bits). The PWMmodulator MOD operates as a D/A converter producing a PWM signal for theoutput circuits PA and NA on the basis of the control code and asign-reversed signal thereof.

The feedback path is as follows. To the voice coil VCM, a resistor Rsfor current sensing is coupled in series. The resistor Rs converts a VCMcurrent Ivcm to a voltage signal. The sense amplifier Ksens senses thevoltage signal converted by the resistor Rs. The output of the senseamplifier is converted to a digital signal by the analog-to-digitalconverter ADC. The ADC is a ΔS ADC. The output signal of the ADC is madethe detection value Ivcmdet through the low-pass filter LPF1, thedecimation filter DFL, and the lag-lead filter LLFL.

For understanding of the present invention, FIG. 14 shows a blockdiagram of a VCM driver using a PWM method investigated by the presentinventors on the basis of Patent Document 2. Details of it are describedin Patent Documents 2 and 3. The forward path of the VCM driver shown inFIG. 14 includes a digital operation type phase compensator, a ΔSmodulator, a PWM modulator MOD, output amplifiers PA and NA, and a voicecoil VCM. The feedback path includes a current sense amplifier Ksens, aΔS type ADC, an LPF1, a decimation filter DFL, and a lag-lead filterLLFL. For improvement of the line transient characteristic and the PSRR,the ADC is provided to detect the power supply voltage Vps and adjustthe gain of the forward path so as not to depend on the power supplyvoltage Vps.

A first point of difference in configuration between the VCM driveraccording to the present invention shown in FIG. 1 and the VCM driver ofFIG. 14 is that the deriving point of a proportional componentcompensation for the PI type phase compensator is changed from a pointsubsequent to error comparison between an input signal IN and thedetection value Ivcmdet of a VCM current Ivcm to a point of Ivcmdetoutput. As a result of this, inside the major loop of FIG. 14(conventional), a minor loop having a first-order characteristicincluding an expanded ΔΣ modulator, a PWM modulator MOD, outputamplifiers PA and NA, a VCM coil, a current sense amplifier Ksens, a ΔStype ADC, an LPF1, a decimation filter DFL, a lag-lead filter LLFL, anda constant Kp can be formed.

FIGS. 2(A) and 2(B) show loop gain characteristic diagrams forexplanation of the present invention. FIG. 2(A) is a loop gaincharacteristic diagram corresponding to the VCM driver of FIG. 14, andFIG. 2(B) is a loop gain characteristic diagram of the presentinvention. As shown in FIG. 2(A), the VCM driver of FIG. 14 uses apole-zero-cancel method and keeps the stability of the system through asingle-pole-operation by canceling the pole of the inductor of the voicecoil by the zero point of the phase compensator. The entire band isdecided with the constant Ki and the zero point is adjusted with Ki/Kp.In other words, a loop gain having a first-order characteristic shownwith a solid line L1 is obtained by selecting ωz (=Ki/Kp) so as to matchthe pole ωl of the inductor shown with a dotted line L2 in FIG. 2(A).

The VCM driver of FIG. 1 uses a minor loop method. The band ωloop1 ofthe minor loop is made flat until at least a target band frequency ωvcmby the function of Kp as shown in FIG. 2(B). In other words, the loopgain of the minor loop is as shown with a dotted line L4, and theclosed-loop gain of the minor loop has a first-order delaycharacteristic having the band ωloop1. In addition, a single poleoperation shown with a dotted line L3 in FIG. 2(B) is obtained by themajor loop performing error comparison between an input signal IN andthe detection value Ivcmdet of a VCM current Ivcm. The entire system isdecided with Kid, and the band ωloop1 of the minor loop is decided withKp.

FIG. 3 is an error suppression characteristic diagram for explanation ofthe present invention. In FIG. 3, the error suppression characteristicof the VCM driver of FIG. 14 is shown with a dotted line, and the errorsuppression characteristic of a VCM driver having a minor loop shown inFIG. 1 is shown with a solid line. In an error of an output voltageexcluding a quantization error which is unique to a digital system,distortions caused by a variation of the power supply voltage Vps anddead times of the output circuits PA and NA are main components, whicharise inside the minor loop. Because of this, the suppressioncharacteristic for the error voltage can be significantly improved fromthe error suppression characteristic of the VCM driver of FIG. 14 by theminor loop method having a second-order characteristic.

The second difference in configuration is that a zero-crossingcompensation circuit PNCNT for the output circuits PA and NA is added.The output circuits PA and NA constitute an H bridge circuit as shown inFIG. 15(A). As shown in FIG. 15(B), a dead time tDEAD is provided ininput PWM signals PWMinP and PWMinN so that an upper arm (MOSFET M1) anda lower arm (MOSFET M2) are not turned on at the same time. This can besimilarly said for the relation between a MOSFET M3 and a MOSFET M4. Forthis reason, the pulse widths of the outputs VCMP and VCMN of the outputamplifiers PA and NA have errors to the PWM pulse width of an input. Theerrors appear as amounts of offset of control, which are reversed whenthe polarity of the coil current Ivcm changes, so that the outputs havea control dead band like the PWM mode characteristic shown with a solidline in FIG. 16 and thereby accurate head position control becomesdifficult.

FIG. 4 is a block diagram of an embodiment of a zero-crossingcompensation circuit according to the present invention. Thezero-crossing compensation circuit compensates the control dead band ofthe PWM mode characteristic. The zero-crossing compensation circuitPNCNT is, so to speak, a first-order Δ-S modulation type errorcorrection circuit. Since the output circuits PA and NA have a deadtime, the pulse widths of the outputs VCMP and VCMN of the outputcircuits PA and NA have errors to the pulse width of an input. For thisreason, the voltages of the output signals VCMP and VCMN are divided byresistors R1 and R2 and resistors R3 and R4, and the polarities of thevoltages are determined by inverter circuits IV1 and IV2, and arecounted using a reference clock MCLK by the polarity counter PNCNT. Thedifference between the counter output and the PWMIN (control code) isadded to an updated control code (PWMIN) in the next cycle to compensatethe zero-crossing error.

Like this, the measuring times of the output pulses (VCMP and VCMN) andthe error time of the indicated value PWMIN of the PWM control are addedto the next input signal of the PWM modulator to correct it after onePWM update cycle delay. As a result of this, the error (zero-crossingdistortion) caused by the dead times of the output amplifiers PA and NAis reduced as shown with a dotted line in FIG. 16. In FIG. 16, a thinsolid line is a linear mode characteristic. As shown in FIG. 16,first-order ΔS type correction by which an error correction amount isadded to an input signal after one PWM update cycle delay is effectivein terms of oscillation, because second-order or higher ΔS typecorrection causes an oscillation operation when the error is large.

The third difference in configuration is that the low-pass filter LPF2and the decimation filter DEC2 are added after the ΔS modulator on theforward path in FIG. 1 to allow the ΔS modulator to operate with ahigher operation clock. In the configuration of FIG. 14, the PWMfrequency and the operation speed (update rate) fds of the ΔS modulatorare limited as fds=fpwm from the viewpoint of the control of occurrenceof an aliasing distortion. By the configuration of the presentinvention, the operation speed of the ΔS modulator can be increased andthe quantization error arising on the forward path can be reduced. Sinceincrease in the operating frequency of the ΔS modulator causes reductionin the decimation ratio of the feedback path, the cut-off frequency ofthe low-pass filter LPF1 on the feedback path can be set high and thedelay time can be thus shortened. In this case, it is feared that thehigh frequency noise increases as the cut-off frequency of the low-passfilter LPF1 on the feedback path increases. However, the configurationof the present invention includes the low-pass filter LPF1 on thefeedback path and the low-pass filter LPF2 on the forward path, so thatthe high frequency noise does not so much increase as compared with theconfiguration (having a low-pass filter LPF1 only on the feedback path)of FIG. 14.

The loop gain of the minor loop has a first-order LPF characteristichaving a breaking point ωl in the coil impedance. The constant Kp isselected appropriately and the band ωloop1 of the minor loop is set at arequired value. The band ωloop1 should usually be selected to be a valuethat is several times higher than the VCM band ωvcm. Furthermore, thehigher the band ωloop1, the more the error suppression characteristic isadvantageous, and the VCM band ωvcm can also be increased, so that theshorter the in-loop delay time of the minor loop, the better it is. Thein-loop delay time can be improved by the third difference inconfiguration.

FIG. 5 is a block diagram of an embodiment of an expanded ΔΣ modulator.In this embodiment, a decimal point appears in the output of asecond-order LPF subsequent to a second-order ΔS modulator, and iscorrected by a decimal separation, a delay stage (Z⁻¹), and an adder. Inthis embodiment, the number of bits (QM) of the ΔS modulator is six, andthe number of bits (M) of the PWM modulator is seven. As shown in FIG.5, this embodiment has an advantage that the hardware can be simplifiedbecause the second-order LPF can be realized using two adding filters.

FIG. 6 is a block diagram of another embodiment of an expanded ΔΣmodulator. In this embodiment, gain adjustment is preliminary performedso that no decimal point appears in the output of a second-order LPF. Inthis embodiment, the number of bits (QM) of the ΔS modulator is six, andthe number of bits (M) of the PWM modulator is seven. Like theembodiment of FIG. 5, this embodiment has an advantage that the hardwarecan be simplified because the second-order LPF can be realized using twoadding filters. In addition, in this embodiment, the hardware can bemore simplified than the embodiment of FIG. 5 because this embodimentcan be realized with only integer operations by performing suitablescaling before the ΔS modulator.

FIGS. 7 to 10 are simulation waveform diagrams for explanation of thepresent invention. FIG. 7 is a waveform diagram corresponding to the VCMdriver of FIG. 14, in which the input signal is equivalent to 2 kHz/10mA. When the input signal has changed from the positive side to thenegative side, a large zero-crossing distortion arises in fallingwaveforms to the negative side of a VCM current and a VCM current afterfiltering. In contrast, when the input signal has changed from thenegative side to the positive side, a large zero-crossing distortionarises in rising waveforms to the positive side of a VCM current and aVCM current after filtering.

FIG. 8 is a waveform diagram in the case that the minor loop in FIG. 1is provided, in which the input signal is equivalent to 2 kHz/10 mA likethe above. Zero-crossing distortions in a VCM current and a VCM currentafter filtering which arise when the input signal has changed from thepositive side to the negative side or when the input signal has changedfrom the negative side to the positive side are significantly improvedas compared with those in FIG. 7 by addition of the minor loop.

FIG. 9 is a waveform diagram in the case that the zero-crossingcompensation circuit in FIG. 1 is provided, in which the input signal isequivalent to 2 kHz/10 mA like the above. Zero-crossing distortions in aVCM current and a VCM current after filtering which arise when the inputsignal has changed from the positive side to the negative side or whenthe input signal has changed from the negative side to the positive sideare significantly improved as compared with those in FIG. 7 by additionof the minor loop.

FIG. 10 is a waveform diagram in the case that the minor loop and thezero-crossing compensation circuit in FIG. 1 are provided, in which theinput signal is equivalent to 2 kHz/10 mA like the above. Zero-crossingdistortions in a VCM current and a VCM current after filtering whicharise when the input signal has changed from the positive side to thenegative side or when the input signal has changed from the negativeside to the positive side are improved as compared with those in FIGS. 8and 9 by addition of the minor loop and the zero-crossing compensationcircuit.

FIG. 11 is a block diagram of another embodiment of a VCM driveraccording to the present invention. In this embodiment, a voice coil VCMis driven linearly. This embodiment is different from the embodimentshown in FIG. 1 in that the expanded ΔΣ modulator and the PWM modulatoron the forward path are replaced with a ΔS type DAC consisting of a ΔSmodulator, a DAC, and an ALPF to drive the voice coil VCM linearly. TheDAC is a digital-to-analog converter. 18 The ALPF is an analog low-passfilter. As a result of this, the quantization error in the PWM modulatorMOD can be reduced, and the zero-crossing distortion can also be reducedby adopting class AB power amplifiers PA and NA in the case of lineardrive, so that the SN ratio can be improved as compared with the VCMdriver shown in FIG. 1.

FIG. 12 is a block diagram of another embodiment of a VCM driveraccording to the present invention. In this embodiment, the PWM methodshown in FIG. 1 and the linear method shown in FIG. 11 are usedalternately. The expanded ΔΣ modulator and the PWM modulator shown inFIG. 1 are almost equivalent to the ΔS type DAC in terms ofconfiguration, and are only different from it in configuration betweenthe output of the ΔS modulator and the inputs of the power amplifiers.Thus, the combination drive method can be realized without a largeincrease in cost. In this configuration, the voice coil is operated bythe PWM method at a seek operation, and is operated by the linear methodat a tracking operation. The signal PWM/LIN is an operation switchingsignal for the output circuits. At switching between both operationmodes, seamless switching between the characteristic (PWM mode withzero-crossing compensation) shown with a dotted line in FIG. 16 and thecharacteristic (linear mode LIN) shown with a thin solid line in FIG. 16is required, so that the entire loop transfer function at PWM(zero-crossing compensation) drive should be matched with that at lineardrive. Also from this viewpoint, LPF2 is needed after the ΔS modulatoron the forward path.

The VCM drivers of FIGS. 1, 11, and 12 are also provided with a serialport for input/output operation between the VCM driver and thecontroller, a circuit of estimating a back electromotive voltage whichcalculates a back electromotive voltage Vb−emf (estimated value) of thevoice coil VCM and supplies it to the controller as speed information,and the like, as described in Patent Documents 2 and 3, although theyare omitted in the VCM drivers shown in FIGS. 1, 11, and 12. It is madepossible to send the back electromotive voltage Vb−emf calculated by thecircuit of estimating a back electromotive voltage to the controllerthrough the serial port. The controller is able to recognize a travelingspeed of the head from the back electromotive voltage received, and useit, for example, for speed control of the voice coil motor performed athead loading to move the magnetic head from the retraction positioncalled a ramp to a position over a disk. If the traveling speed of themagnetic head is too high, the magnetic head may come into contact withthe surface of the disk to damage the disk. However, it can be preventedby the speed control.

FIG. 13 is a schematic configuration diagram of an embodiment of amagnetic disk device to which the present invention is applied. In thehard disk drive (HDD), data is written onto and read from a disk 2 beingrotated at a high speed by a spindle motor 1 through a head 3. The HDDuses a voice coil motor (VCM) 4 which is a head actuator changing astorage position (position of the head 3), and performs a feedbackcontrol of reading servo information previously stored on the disk 2with a signal processing IC 5, issuing a drive current command fordriving the VCM 4 with a controller 6 including a microcomputer, anddriving the VCM 4 with a digital control type VCM driver 7 shown in FIG.1, 11, or 12. The VCM driver 7 itself, or the VCM driver 7 and othercircuits such as a drive control circuit (not shown) of the spindlemotor 1, are composed of one semiconductor integrated circuit device. Byapplying the present invention to a HDD, it can be realized to increasethe speeds of the seek operation and tracking operation of the HDD andreduce the power consumption of the HDD.

FIG. 17 is a characteristic diagram showing the relation between the PWMfrequency and the quantization error. FIG. 17 shows the relation in thecase of second-order ΔS modulation. When the PWM frequency is reduced byone-half in the case of the same clock CLK (50 MHz), the noise isimproved by “−15 dB (ΔS modulator)+6 dB (PWM)=−9 dB”. M is the number ofbits of the modulator MOD. From the relation of “PWM frequencyfpwm=CLK/2^(M)”, when M is increased to, for example, nine, the PWMfrequency fpwm decreases and the update rate increases, so that eachnoise amplitude increases as shown in FIG. 17. Increase of the noiseamplitude causes the deterioration of the accuracy of the positioncontrol of the magnetic head. When the frequency of the input signal INis set to 50 KHz like the above embodiment, the noise amplitude matchesthe target value shown in FIG. 17 when M is the order of 7 inconsideration of noise reduction caused by a feedback loop such as amajor loop or a minor loop.

A zero-crossing distortion in the PWM method is a duty error of PWMcaused by a dead time, and can be thus reduced by actually measuring andfeeding back an error time like the embodiment of FIG. 4. In ahigh-order ΔS modulator, the degree of S/N improvement associated withthe rise in the operating frequency is large. For example, in asecond-order ΔS modulator, when the operating frequency rises by twice,the S/N can be improved by 15 dB. Similarly in a multi-bit PWM methodusing ΔS modulation, the decrement of noise by the increase in the speedof the ΔS modulation is larger than the increment of noise by thereduction in the number of bits of the PWM modulator as shown in FIG.17. Furthermore, in general, when the PWM frequency rises, the switchingloss increases. In other words, the PWM frequency and the noise are in atrade-off relation, and the selection of a PWM frequency is needed. Inthis case, it is desirable that the characteristic is not changed andonly the PWM frequency is changed on the feedback path and up to the ΔSmodulator on the forward path.

However, in an operation speed of ΔS modulation higher than the PWMfrequency, the noise increases due to an aliasing distortion, so that alow-pass filter (LPF) and a decimation filter which are suitable for theratio between the PWM frequency and the speed of the ΔS modulator areadded after the ΔS modulator in order to prevent the noise increase. Inthe case of the above embodiment, the feedback path and the forward pathhave LPF1 and LPF2, respectively, so that it becomes possible todistribute orders to LPF1 and LPF2, a high-order LPF having a shortdelay time can be realized, the delay time of the loop can be optimized,and the S/N of ΔS modulation of the forward path can also be optimized.Furthermore, when the delay time of the LPF becomes short, the controlband can be made wider as compared with the configuration of FIG. 14,and a wide band minor loop feedback can be introduced. In this case, anegative feedback effect of the minor loop and a negative feedbackeffect of the major loop can be obtained at the same time, so that asecond-order error suppression characteristic is obtained as a result,and thereby both of an error and a zero-crossing distortion at an outputstage caused by a power supply variation can be reduced. As a result ofthis, the ADC for correcting the error at the output stage due to thepower supply variation can be omitted.

By preparing a path for replacing PWM drive with linear drive like theembodiment of FIG. 11, a noise caused by a quantization error arising atthe PWM drive can be reduced. Furthermore, by using a PWM operation anda linear drive path in combination like the embodiment of FIG. 12, itbecomes possible to use a power consumption priority operation and anoise priority operation properly, and further improvement of the systemperformance becomes possible. For example, in the case of a HDD, byusing the linear drive at the seeking in which power consumptionreduction is important and using the linear drive at the tracking inwhich accuracy is important, low power consumption and high accuracy VCMdrive control becomes possible. In this case, by separating the pathbetween the output of the ΔS modulator and the input of the outputcircuits PA and NA of the forward path to a linear drive and a PWMcontrol, the cost can be minimized.

A circuit having the configuration shown in FIG. 4 can be widely used asa PWM amplifier which receives a digital value input signal and producesan analog output signal corresponding to the input signal. The circuitcan also be used as an audio digital amplifier compensating an outputdistortion at a dead time in addition to a PWM modulator of a VCM driverlike the above embodiment.

Up to this point, the present invention developed by the presentinventors has been concretely described based on the embodiments.However, the present invention is not limited to the embodiments andvarious changes and modifications can be made without departing from thespirit and scope of the present invention. For example, when thecharacteristics of the sense amplifier are used alternately, there is acase that a variation at switching becomes a problem by the delay timeof a low-pass filter on the feedback path. In this case, LPF1, DFL, andLLFL after the ΔS ADC on the feedback path in FIG. 1 may be deleted asappropriate. For example, concrete configurations of concrete circuitsrealizing the circuit blocks, amplifiers constituting the outputcircuits PA and NA, and switching circuits added as shown in FIG. 12 maybe realized in various embodiments. Furthermore, a voltage negativefeedback type circuit which performs negative feedback of both or eitherof the output voltages during PWM operation to produce a drive voltagemay also be used.

The present invention may be widely applied to a VCM driver such as aHDD or the like and a PWM amplifier operated by a digital input signal.

1. A VCM driver comprising a forward path and a feedback path, therebyproducing a drive current of a voice coil motor for position control ofa magnetic head using as an input signal a digital current controlsignal being formed corresponding to positional information read fromthe magnetic head and position command information from a controller,wherein the forward path includes: a digital operation type phasecompensator receiving the input signal; a ΔΣ modulator receiving anoutput signal of the phase compensator to convert the output signal intoa control code of predetermined bits; a PWM modulator receiving thecontrol code converted by the ΔΣ modulator to produce a PWM signal; andan output circuit receiving the PWM signal produced by the PWM modulatorto drive a voice coil; wherein the feedback path includes: a senseamplifier sensing a current flowing in the voice coil; an ADC receivingan output signal of the sense amplifier; a first low-pass filterreceiving an output signal of the ADC; and a first decimation filterreceiving an output signal of the first low-pass filter; and wherein anoutput signal of the decimation filter is fed back to the input side ofthe phase compensator to form a major feedback loop having a first-ordercharacteristic loop gain; and an output signal of the decimation filteris fed back to the output side of the phase compensator to form a minorfeedback loop having a loop gain being flat until a target bandfrequency when viewed from the output side of the phase compensator. 2.The VCM driver according to claim 1, wherein the PWM modulator counts areference clock to produce the PWM signal having a duty ratiocorresponding to the control code.
 3. The VCM driver according to claim1, wherein the forward path further includes a second low-pass filterand a second decimation filter after the ΔΣ modulator.
 4. The VCM driveraccording to claim 3, wherein a selection circuit is provided in thesecond low-pass filter and the second decimation filter, using thecontrol code converted by the ΔΣ modulator as an input signal having thenumber of bits adapted to a PWM frequency of the PWM modulator.
 5. TheVCM driver according to claim 1, further comprising a zero-crossingcompensation circuit which detects an error pulse between the PWM signalinput to the output circuit and an output signal of the output circuitcorresponding to the PWM signal and adds the error pulse to a PWM signalto be input next.
 6. The VCM driver according to claim 5, wherein thezero-crossing compensation circuit includes: a dead time generatingcircuit of the output circuit; a counter counting a reference clock toproduce a pulse corresponding to an error time corresponding to anactual pulse width of an output amplifier produced by the dead timegenerating circuit and a pulse width of the input PWM pulse; and anaddition circuit adding the pulse produced by the counter to a PWM pulseto be input next.
 7. The VCM driver according to claim 1, furthercomprising: a DAC converting an output code of the ΔΣ modulator into ananalog signal; and a third low-pass filter receiving an output signal ofthe DAC to produce a drive control signal corresponding to the PWMsignal, instead of the PWM modulator.
 8. The VCM driver according toclaim 1, further comprising: a first driving unit which passes an outputcode of ΔΣ through a second low-pass filter and a second decimationfilter and then inputs an output of the second decimation filter to thePWM modulator to produce a PWM signal for driving an output amplifier;and a second driving unit which passes the output code of the ΔΣmodulator through the DAC converting the output code to an analog signaland then inputs an output signal of the DAC to a third low-pass filterto produce a linear signal for driving the output amplifier, wherein theoutput circuit is able to be switched between a first operation mode ofselecting the first driving unit and a second operation mode ofselecting the second driving unit; the first operation mode is set whena seek operation is performed; and the second operation mode is set whena tracking operation is performed.
 9. The VCM driver according to claim8, further comprising a zero-crossing compensation circuit which detectsan error pulse between a PWM signal input to the output circuit and anoutput signal of the output circuit corresponding to the PWM signal andadds the error pulse to a PWM signal to be input next in the first mode.